Mixer circuits are widely employed in radio frequency (‘RF’) communication systems. The expression ‘radio frequency’ is used in this specification to designate wireless communication frequencies without any specific upper limit and embodiments of this invention are usable up to millimetric wavelength frequencies and beyond.
Modern wireless communication systems have stringent dynamic range requirements. The dynamic range of a receiver, in particular, is often limited by the first down conversion mixer.
Generally speaking, mixers perform frequency translation by multiplying two signals (and possibly their harmonics). Down conversion mixers employed in the receive path have two inputs, the RF signal to be down converted and a waveform at a selected frequency generated by a local oscillator (‘LO’), these signals being applied to an RF port and an LO port of the mixer respectively.
The performance parameters of typical down conversion mixers are compromises between parameters such as the noise figure, the linearity, the conversion gain, the input impedance, the 3rd order intercept point and the port-to-port isolation. The noise figure is important in mixers as it is a measure of how much noise the mixer adds in the system. The input impedance of the receiver should be well matched to increase the conversion gain. The 3rd order intercept point is important as it is a measure for the linearity of the mixer. The port to port isolation is also an important issue, since the LO-RF feed-through results in LO leakage to the low noise amplifier of the receiver and eventually the antenna, whereas the RF-LO feed-through allows strong interferers in the RF path to interact with the local oscillator driving the mixer. The feed-through from the LO to the intermediate frequency (‘IF’) output of the mixer, LO-IF, is important because if substantial LO signal exists at the IF output even after low-pass filtering, then the following stage may be desensitized. Finally, the RF-IF isolation determines what fraction of the signal in the RF path directly appears in the IF.
FIG. 1 shows the main elements of a known double-balanced Gilbert mixer 100, including an RF stage and an LO stage connected in series between voltage supply rails +V and −V. The RF stage receives the RF signal at a port comprising terminals RF+ and RF− and the LO stage receives the LO signal at a port comprising terminals LO+ and LO−. The output port comprises terminals IF+ and IF− connected to interconnections 108 and 110 between the LO stage and respective output impedances ZL connected to the supply rail +V. The RF stage comprises a differential pair 102 of bipolar transistors in emitter follower configuration with a current source. The LO stage comprises double differential pairs 104 and 106 of bipolar transistors cross-coupled to steer the current from one side to the other side of the differential pairs by commutating alternately all the tail current in the RF stage and the current source from one side to the other of the LO stage at the LO frequency. If the input signals are single-ended they can be applied to the input ports of the Gilbert cell via an input matched balanced-unbalanced transformer (‘balun’). Other Gilbert cell mixers are known using field effect transistors, such as complementary metal-oxide field effect transistors, instead of bipolar transistors.
The article “Design Considerations for Low-Noise, Highly-Linear Millimeter-Wave Mixers in SiGe Bipolar Technology”, by S. Trotta et al. in IEEE ESSCIRC 2007 Digest describes a mixer circuit 200 derived from the Gilbert mixer and is shown in FIG. 2. The RF stage comprises transistors T1 and the LO stage comprises transistors T2 and T3. Notably, the circuit uses transmission lines L3 of length λ/4, where λ is the wavelength of the RF signal involved, so that they present an inductive impedance to provide the bias voltage VbRF to the bases of the RF differential pair T1 without attenuating the RF signal. The second harmonics at 2ωRF at node A in the LO stage are strongly attenuated by λ/4 transmission lines L1, which transform the low impedances at the emitters of the differential pairs T2 and T3 (node A) to high impedance at the collectors of the RF differential pair T1, the second harmonics at 2ωRF being grounded through the low impedance path including L3, which is a λ/2 transmission line at 2ωRF. In this mixer circuit, L1 improves the linearity and noise figure compared to the circuit of FIG. 1. However, the bias voltages for the RF pair and the LO pairs are not independent, which degrades the noise figure. Also a large voltage supply is needed.
U.S. Pat. No. 6,232,848 describes various circuits, including a mixer circuit 300 shown in present FIG. 3 and having an RF stage comprising a differential pair 102 of bipolar transistors and an LO stage comprising double differential pairs 104, 106 of bipolar transistors, with coupling elements providing parallel DC connections for the RF and LO stages respectively between the DC voltage supply rails. Resonant elements ZL, CL1, CL2, ZL, CL3, CL4 in the DC connections are coupled through DC isolating capacitors CC to provide a series RF connection for the RF and LO stages between the supply rails so as to produce a mixed differential amplified signal at differential output terminals connected with the amplifier paths of the LO stage. The resonant elements are tank circuits which provide low resistance paths for the DC supply current with the supply rails +V and −V, thereby avoiding applying the DC voltage drop across the RF stage current source 112 to the LO stage. At the resonance frequency, the tank circuit exhibits high impedance to ground, mimicking an AC current source. However, this circuit configuration presents several problems related to the biasing, imbalance, mismatch rejection, noise, temperature compensation and stability, especially of the RF stage.